System for creating and amplifying three dimensional sound employing phase distribution and duty cycle modulation of a high frequency digital signal

ABSTRACT

A method of presenting audio information where changes in amplitude and changes in frequency in two channels (stereo) have the additional parameter of phase information added to re-create the feeling of a live performance. Also, all three parameters are converted into duty cycle modulation of a high frequency digital pulse. Conventional loudspeakers and the brain decode the signal to provide audio signals that contain more information than simply frequency and amplitude changes as a function of time.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a division of co-pending application Ser. No.08/887,303, filed Jul. 2, 1997.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to audio signal processing and, moreparticularly, to a signal processor that produces an effect thatresembles a live performance, restoring what was lost during thetransduction and recording sequences.

2. Description of the Background

Some audio processing systems make use of amplitude or frequencyadjustment or both, others rely on optimizing the Group Delays offrequencies; however, the current invention described herein convertsaudio information (changes in amplitude and frequency) into a phasespace, where changes in amplitude and frequency become phase shifts ofportions of a digital pulse. Regular loudspeakers can automaticallydecode the phase information, resulting in a virtually distortion-freeanalog acoustic signal whose bandwidth is approximately 0-50 KHz andwhose characteristics include not only changes in amplitude andfrequency, but, also, changes in phase as a function of frequency. It isobvious from the scientific literature that no coherent theory ofhearing currently exists. Low frequency acoustic signals' wavelengthsare much too large to enter the ear canal, and high frequencies (above 6kHz) cause bifurcation of neuronal firing in the brain, allowing forsubconscious processing of frequencies considered outside the (20-20kHz) hearing range. The current invention expands the bandwidth of theaudio signal from the “normal” frequency range to about 0-50 KHz,separates frequencies as a function of time, and converts ordinarystereo signals into phase distributed monaural. The result is a signalthat interacts with the human brain in a new way to produce an effectthat resembles a live performance, restoring what was lost during thetransduction and recording sequences.

It was discovered in 1932 that hearing was not strictly physical, thatpsychological factors also contributed to our perception of sound (OnMinimum Audible Sound Fields, by Sivian and White. Presented to theAcoustical Soc. of Am. at Ann Arbor, Mich. Nov. 29, 1932). That phaseangles of even pure tones are sensed by humans was established in 1956(Just Noticeable Differences in Dichotic Phase, by Zwislocki andFeldman, in J. Acous. Soc. of Am., Vol. 28, #5, September 1956). The earis a non-linear device—current models and theories of hearing are based,primarily, on old, outdated linear models and concepts (AuditoryFrequency Selectivity, edited by Moore and Patterson, NATO ASI series A,Vol 119, 1986). Some musical harmonics are non-linear (violin overtones,e.g.) (Regarding the Sound Quality of Violins and a Scientific Basis forViolin Construction, by H. Meinel, Ln, J. Acous. Soc. Am., Vol 29, #7,July 1957). The interaction of acoustic signals from various musicalinstruments (including Human voices and electronic synthesizers) createinterference patterns that are embedded in the recording medium (tape,e.g.), but whose characteristics are ignored with current transducersand recording and playback equipment (Both Sides of the Mirror:Integrating Physics and Acoustics with Personal Experience, by HelenHall, in, Leonardo Music Journal, vol 3, pp17-23, 1993). Just as laserprocessing of images focused on photographic emulsions can bring out thethree-dimensional information in the two-dimensional image by retrievingphase information in the interference patterns, so this inventionrestores 3D information lost during transduction and recording. Theresult is a restoration of the “live” performance.

Classical theory indicates that acoustic events can be described in atleast two ways; in a time domain or a frequency domain, each convertibleinto the other via a Fourier transformation. The mathematicalformulation for this process is well known. The time-domaincharacterization of an acoustical event is a scalar, while thefrequency-domain representation is a complex vector quantity containingamplitude and phase information. The time domain representation can alsobe expressed as a complex quantity. The scalar portion of the timedomain vector represents performance based on impulse excitation; theimaginary part of the vector is the Hilbert transform of the scalar.

Loudspeakers and electrical networks which transfer energy from one formto another can be characterized by response to an impulse function,because the impulse response can be manipulated to predict the behaviorof the system in response to any arbitrary signal. Fourier transformswork for predictive systems as well as causal systems. However, thegroup velocity of a set of audio signals is not related to time delayfor all possible systems, and uniform group delay does not insurb adistortionless system. Group Delay is derived from phase delay, which isdefined as the phase shift in a system at a given frequency. Group delayis associated with a group of frequencies around a central carrier, suchas those encountered in modulated communications, but it also finds somerelevance in describing how a system responds to a change in frequency.Group Delay follows an inverse square law. The value is fixed for DC butit approaches a finite value (near zero) at infinity. For a givenfunction, and given the appropriate values for the resistor andcapacitor, this logarithmic response will appear across the audio range.Starting with Tgd=2α₀/α₀ ²+ω², it can be shown that:

Tgd(ω≈α₀)≈2.3/α₀log(α₀/ω)

For a simple case it is possible to relate a logarithmic approximationto the group delay. The approximation was developed around a regionwhere alpha equals omega. A more general equation for a larger region ispresented below. It was derived using similar techniques but spans thearea from omega equals alpha to omega “large” (50K radians or so). Smallvalues of omega are not permissible, and the error at omega equals alphais significant. These logarithmic equations are not specificallynecessary for the design process but when the user works with practicalcircuits, it will be noted (on test equipment) that the Group Delay ofthe audio bandwidth changes logarithmically with frequency. Thefollowing equation can be used to validate the observations; however, itis noted that because of the foregoing, Group Delay is rathermeaningless and phase shift more accurately describes the true action ofthe circuit. Group Delay is included here to provide an alternate way ofanalyzing the circuit's action.

These two equations are generally equivalent for ω>5α₀:

Tgd(ω)=2α₀/α₀ ²+ω²

Tgd(ω)=2α₀*ln[1+(α₀/ω)]²

The same equation rewritten for standard logarithms:

Tgd(ω)=4.6/α₀*log[1+(α₀/ω)]²

Interaural time difference, necessary for determining the position of asource, also has bearing on pitch. Further, most of the criticalinteraural time differences are in the range of plus or minus 1millisecond. Thus, when the group delay is modified, so is theperception of the sound.

A generalized version of the All-pass response Group Delay is presentedbelow. This equation can be used, with reasonable accuracy, to predictthe group delay of a specific frequency for various RC combinations. Italso accounts for Gain adjustments. Using these equations, one cantailor the Group Delay response. Referring to FIG. 5:

α₀=1/R ₁ C and A=R ₃ /R ₂.

The general transfer function is:

T(s)=−As−α ₀ /s+α ₀

which means the gain of the circuit is:

|T(s)|=−A

The phase response is:

φ(ω)=−2tan⁻¹(ω{square root over (A)}/α ₀)

and the Group Delay is given by:

Tgd(ω)=(A+1)α₀/α₀ ²+ω_(A) ² +A*50 ns+100 ns

The second and third terms are included because their exclusion yieldsincreasingly poor results with increasing frequencies. The aboveequations may be interpreted in the following physical sense: alphadetermines the range over which the group delay respondslogarithmically. An increase in alpha will tend to shift the range tohigher frequencies, but will reduce the group delay itself, i.e., theactual delay times will decrease. A decrease in Gain, A, can be used tooffset this decrease in delay. Conversely, for a given alpha, adjustingGain can be used to set the delay time at the expense of the overallfrequency range. Increasing Gain increases the maximum delay presentedby the system (at very low omega), but the delay of a specific frequencycompared to unity gain will be smaller due to the shift in frequencyrange; adjusting alpha can be used to compensate.

The circuits shown in FIGS. 3 & 5 all utilize an alpha of about 100radians each. Increasing alpha will tend to emphasize lower frequencies,and decreasing alpha will tend to emphasize higher frequencies. In anycase, a specifically desired response will require the adjustment ofboth Gain and alpha.

FIG. 3 shows the cascaded implementation. The effect of the cascade is alinear addition of the delays. The general effect of cascading is todelay a broader range of frequencies by a greater amount, thus enhancingthe effect.

Because the time and frequency domains are two ways of describing thesame event, accurate time domain representation cannot be obtained fromlimited frequency domain information. For example, the time delay of afrequency component passing through a system with non-uniform responsecannot be determined with accuracy. However, a joint time-frequencycharacterization can be made using first and second order all-passnetworks. This is consistent with ordinary human experience. At anyfrequency there are multiple arrivals of the audio signal at thelistener's location as a function of time.

The individual time-frequency components of an audio signal, predictedmathematically, overlap in the time and frequency domains. Therefore, agraphical presentation is not possible, because it is impossible toseparate simultaneous arrival times in a single time domain Plot.

Potential energy (i.e., pressure expressed in dB) and comparisons ofinput to output signals directly (a measure of distortion) do notcompletely describe the performance of audio equipment quality such asloudspeakers, microphone, and electrical networks. Total sound energyprovides phase distortion information and, although phase is notdetectable consciously for simple signals, there are indications thatthe human hearing mechanism is capable of processing complex functionsand perceiving phase information as part of total sound perception.

The square root of the total energy density vector E is equal to the sumof the square root of the potential energy vector and the imaginarycomponent of the square root of the kinetic energy vector:

{square root over (E)}={square root over (P)}+i{square root over (K)}

Attempts to measure the total energy density at a microphone respondingto a remote sound source will only yield part of the total energydensity of the source. Thus, at any given moment, a microphone will notdirectly measure E. Essentially, a microphone compresses complexspatial, multi-dimensional acoustic signals into a single point in timeand space, effectively making the signal two-dimensional as a functionof time. However, the information necessary to unravel the entireoriginal signal is contained in the compressed signal and can beretrieved if processed property.

Although the threshold of hearing has been established in terms ofvector orientation and frequency of pure tones (see, Sivian and S.White, supra), pure tones have no Fourier transforms. The human hearingmechanism processes total energy density, not just the “minimum audiblepressure” associated with a pure audio tone.

The ability to localize direction and distance from a sound source hassomething to do with the orientation of the ear with respect to thevector components of sound. For pure tones, simply the phase differencesbetween arrival of the signal at the two ears provides a clue to thedirection of the source. See, Kinsler and Frey, Fundamentals ofAcoustics (New York: John Wiley and Sons, 1950), pp. 370-392. Thus, theminimum audible field for binaural hearing varies with amplitude,frequency, and azimuth relative to the source signal.

J. Zwislocki and R. Feldman (1956) “Just Noticeable Differences inDichotic Phase”, J. Acoust. Soc. Am., Vol. 28, No. 5, p. 860 (Spetember1956) pointed out that the ears may not be able to detect phase or timedifferences abuve 1300 Hertz and the only directional Qlueo aboye 1300Hz are contained in relative intensity differences at the ears.

In reality, the human auditory system binaurally localizes sounds incomplex, spherical, three dimensional space using two sensors (ears)that are unlike microphones, a computer (brain) that is unlike anycomputer constructed by man, and, at a live performance, the eyes. Theeyes allow us to “hear” direction by providing a sensory adjunct to theears for localization of sound in azimuth, distance and height. Duringreconstruction of a familiar sound, such as a symphony orchestra, thebrain remembers instrument placement and correlates this informationwith auditory clues to provide a more complete sense of the individualorchestra sections and sometimes of the locations of individualinstruments. Techniques for localizing sound direction by the ears,neural pathways, and the brain have been termed “psychoacoustics”.

In addition to direction, the brain will interpret distance as afunction of intensity and time of arrival differences. These clues canbe provided by reflected sound in a closed environment such as a concerthall, or by other means for sound originating in environments where noreflections occur, such as in a large open field. In a closedenvironment, there is a damping effect as a function of frequency due toreverberations. When acoustic energy is reflected from a surface, aportion of the energy is lost in the form of heat. Low frequencies tendto lose less energy and are transmitted more readily, whereas highfrequencies tend to be absorbed more quickly. This makes the decay tipeof high frequencies shorter than that of low frequencies. The air itselfabsorbs all frequencies, with greater absorbtion occurring at highfrequencies.

In “Biophysical Basis of Sound Communication” by A. Michelson (in B.Lewis (ed.), Bioacoustics, A Comparative Approach (London: AcademicPress 1983), pages 21-22, the absorption of sound in air is described asa combination of dissipation due to heat and other factors not wellunderstood. In air, the absorption coefficient in dB/100 meters is 1 atabout 2 khz. At about 9 khz, the signal is down by 10 dB; at 20 khz itis down by 100 dB; and at 100 khz (the upper harmonics of a cymbalcrash), it is down by about 1000 dB. Thus, higher harmonics generated bymusical instruments are drastically attenuated (in a logarithmicfashion) by even a distance of a few feet when traveling to microphones,and then even more when traveling from speakers to the listener's ears.

With conventional stereophonic sound reproduction systems, it isnecessary to be equidistant from the speakers in order to experience theproper stereo effect. With earphones, standard stereo provides a strangepingpong effect coupled with an elevated “center stage” in the middleand slightly above the head. At best, ordinary stereo is an attempt tospread sound out for increased realism, but it is still basicallytwo-dimensional.

In the 1920s Sir Oliver Lodge tested human hearing range out to 100 khz.It has been suggested that the true range of human hearing is notcompletely known. However, the outer ear, inner ear (cochlea), auditorynerve, and human brain are capable of detecting, routing, and processingfrequencies in excess of 100 khz, and possibly to 300 khz and beyond.However, conscious hearing is limited by the brain to roughly 20 hz to20 khz.

There is no currently accepted theory of how humans actually hearoutside the voice range of acoustic signals.

Below about 20 Hz, the wavelength of an acoustic pressure wave is toolarge to enter the ear canal. Experience with low frequency standingwaves suggests an interaction with the cochlea or auditory nervedirectly. Indeed, standing wave acoustic emitters produce the perceptionof distortion-free sound throughout the hearing range. Above about 6 Hz,the “volley” theory and active cochlear processes could account for anincrease in hearing range beyond 20 khz. The volley theory is derivedfrom the fact that there is not a single stimulus-response event pernerve; rather, higher frequency stimulation results in a multiplicity ofneural firings. The process is one of bifurcation wherein the higherfrequencies cause a greater number of neurons to fire. This suggests thepossibility of fractal pattern generation. How the brain interprets thevolley of information presented to it is unknown, however.

In “Auditory Function”, edited by G. Edleman, W. Gall, and W. Cowan,(New York: John Wiley & Sons, 1986), a class of experiments is describedwhich demonstrate acoustic emissions from animal and human ears. Thecochlea can function as a generator of acoustic signals which cancombine with incoming signals to produce higher frequencies. Bothempirical and theoretical studies (indicating that active cochleaprocesses are necessary for basilar membrane tuning properties) supportthe concept.

P. Zurek, in “Acoustic Emissions from the Ear—A Summary of Results fromHumans and Animals”, J. Acoust. Soc. Am., Vol. 78, No. 1, pp. 340-344(July 1985), indicates that frequency selectivity results from activecochlear processes. When the ear is presented with a nonlinear pulse, inaddition to the stimulus response mechanism, another response with an 8millisecond (or longer) delay is produced. This,phase-shifted signal,generated by the ear, may play a role in the actual way in which we hearmusic and other high frequency sounds. When musical instruments producesound, the various Fourier waveforms are not simply producedindependently of each other, but exist in a phase space wherein thereare phase interactions among all of the sounds. Even a single stringplucked on a harp or struck on a piano will produce phase-relatedsignals and harmonics, not simply frequencies and amplitudes. Thus, theear must be capable of decoding phase information in order to properlytransduce complex sounds such as music.

The evoked time-delayed response in the ear is not simply aphase-shifted replica of the original sound, because the higherfrequency components are time delayed less (about 8-10 milliseconds)than the lower frequency components of the emission (about 10-15milliseconds) Also, the amplitude of the evoked response is non-linearwith respect to the stimulus for high stimulus levels, amounting toabout 1 dB for every 3 dB increase in the stimulus. The interaction ofthe stimulus and acoustic emission occurs increasingly with lower andlower levels of input, suggesting that the ear may have a compensationmechanism for low level signals. People with certain types of hearingloss do not product acoustic emissions. At low levels of auditorystimulus, the emissions are almost equal in amplitude to the incomingsignal itself, and they occur even for pure tones. The ear can generatecontinuous signals, and generated signals as high as 8 khz have beenobserved.

As noted earlier, the conscious hearing range is roughly between 20 hzand 20 khz. Audio equipment has been designed to be optimal within thatrange. Also, most equipment has been designed to accurately reproducethat which has been transduced and recorded. However, live sound is atransient phenomenon. It is not possible to compare a live sound withanything, because in order to do so, it must be transduced and recordedin some way. It is this fact that forms the motivation for the presentinvention, and discussion of the prior art that follows.

There have been many attempts to unravel the compressed information inrecorded sound to provide the information that was present in the liveperformance. Most of these attempts have colored the sound and failedbecause our understanding of how we hear has yet to be determined.However, progress has been made, and new theories have pointed the waytoward a path that provides answers to previous questions concerningexactly how the human ear and brain interprets audio information.

Byrd in 1990 (PCT/US91/09375) described a circuit for adding phaseinformation to audio signals in order to restore information that waslost during the transduction and recording process.

Tominari (1989) described a phase shift network that delayed lowfrequencies in time to provide an echo.

Other attempts, although different than Tominari's, suffered from thesame basic problem: how to restore the feeling of the live performancewithout causing some unwanted side effects. Even Byrd's design sufferedfrom loss of a “center” such that vocals seemed to be in a tunnel,although instrumental music came alive with no side effects (there is nocreated center instrumental recordings).

Visser (1985) cites 35 US and Foreign Patents that had attempted tocreate sound in various new ways. He considered his idea to be betterthan any of them, yet his addition of high-frequency broadband noise toaudio signals and to transform the resultant to a binary codedpulse-width-modulation (while useful for some applications) is anunnecessary complication for purposes of creating a phase space out ofan analog signal. The current invention overcomes all shortcomings ofall the prior art and produces not only a re-creation of the liveperformance, but also provides a means for converting the processedsignals into distortion-free digital duty-cyclemodulation and amplifyingthe result to virtually any desired power level at the most efficientand lowest cost possible.

Disclosure Document #344/171 describes in block diagram form a circuitthat was reduced to practice on Nov. 19, 1992. Units were loanedunder-nondisclosure agreements for evaluation, and when it was apparentthat significant advance in sound processing had occurred, DisclosureDocument #374/894 was filed on Apr. 24, 1995 detailing the circuitsinvolved. The PWM chip that converts the output of the device fromanalog to digital and the amplifier circuit was added in July 1995. Thecircuits described herein represent a significant and unobviousimprovement to the circuits described in PCT US91/09375.

Although the prior art has attempted to correct some of the problemsassociated with distortion in audio systems due to phase shifts as afunction of frequency, and spatial distortion due to the inherentinaccuracies in standard stereo, these attempts have not completelysucceeded in restoring lost realism to recorded sound. At best, someprior art processors create the illusion of ambience.

The prior art provides single and, in some cases, multiple correctionsto recorded signals. The object of the prior art is, in general, tocontrol the location of sound cues and provide phase correction, toincrease the quality of the sound by putting back in to the signal whatwas removed by the transduction, recording, and playback systems.

As previously pointed out, microphones compress signals that can consistof many fundamental frequencies from different instruments at differentspatial locations. These signals also contain complex interactions forthe fundamentals and harmonics produced by the same instruments. Whencymbals crash, for example, the harmonics produced reach above 100,000Hertz. As the complex signal develops from these interactions, it canbecome non-linear and sub-harmonics will be present.

At first, it would appear impossible to retrieve or reconstruct acomplex signal whose spectral content has been compressed by microphonesin both the time and spatial domains. The digital sampling rate ofinformation that is recorded on compact discs and digital audio tapes,for example, results not only in a loss of information, but also in anabsolute frequency cutoff that is lower than the upper harmonicsproduced by some musical instruments. The present invention arises fromthe recognition that, if the harmonics and subharmonics of recordedsound are allowed to develop from the fundamental frequencies, and ifthe spectral content of the signal is spatially separated, the originallive sound can be recreated and converted into a digital signal thatcontains all necessary information for the ear and brain to interpretand recreate the original live sound.

SUMMARY OF THE INVENTION

It is, therefore, an object of the present invention to cause harmonicsand sub-harmonics to develop for all frequencies, by continuouslycorrecting the phase of the signal logarithmically as a function offrequency, by spatially separating the spectral content of the signal,by increasing the audio bandwidth of the signal, and digitizing theresult.

The invention is based, in part, on the recognition that the humanhearing mechanism for sensing audio signals (in contrast toelectromagnetic and tactile signals) is different from the electroniccircuits used to construct amplifiers, microphones, tape recorders, andother types of audio equipment. Thus, when humans hear or sense an audiosignal, it is processed differently than standard apparatus attemptingto transduce, record, and playback the original signals.

The present invention provides a new way to process and amplify sound ina way that converts amplitude, frequency, and phase information intoduty cycle modulation of a high frequency digital pulse. The signal isintegrated by the voice coil of ordinary loudspeakers and the phaseinformation is interpreted by the brain so as to provide threedimensional sound. The acoustic hologram so produced is perceived to belike a live performance. Simple digital switching amplifiers can beadded to yield any desired power level.

The invention has several objects:

to create an expanded bandwidth for recorded (and live) sound in orderto take advantage of harmonics outside the “normal” (20-20 khz) hearingrange;

to create a phase shift of frequencies such that higher frequencieseffectively reach the ear after lower frequencies (this creates thethree dimensional characteristics of the sounds);

to allow natural harmonics to be generated (this provides a sense ofbeing closer to the source);

to convert the amplitude, phase, and frequency information into dutycycle modulation of a high frequency digital pulse (>43 KHz) in order toencode the information in a way the ear and loudspeaker can preciselyrecreate the original information;

and to amplify the result with a low distortion, simple, inexpensiveamplifier that has no feedback.

In accordance with one aspect of the present invention, an audio signalprocessor is provided comprising an input terminal for receiving anaudio signal, first, second, and third processing stages for processingthe audio signal, and an-output terminal for coupling the processedaudio signal to an output device. The first and second signal processingstages are arranged in a series or cascade configuration, and each stagefunctions to phase shift fundamental and harmonic frequencies as afunction of frequency. The phase shift increases in a negative directionwith increasing frequency, so that higher frequency signals lag thelower frequency signals. Also, the left and right channels are crossedover twice in order to homogenize the signal into phase distributedmonaural. The output is then fed into a digital chip that converts theamplitude, frequency, and phase information into a form of duty cyclemodulation.

The present invention is implemented by means of a relatively simpleelectronic circuit that can be manufactured and sold at very low cost.The principal components of the circuit can, if desired, be reduced to asingle dual inline package (DIP) which can be incorporated into existingtypes of audio equipment. The invention can be utilized with nearly allexisting types of power amplifiers, stereo tuners, and phonographs withpreamplifiers, as well as with compact disk (CD) players, digital audiotape (DAT) players, and conventional analog tape recorders and players.All recorded media can be reproduced with a sound that is close to thatof a live performance.

The invention can be used with any number of audio channels or speakers;the resulting sound will be dimensionalized, to some extent, with even asingle speaker. The signal processing that is carried out by the presentinvention transfers to tape and to virtually any other type of recordingmedium. Thus, for example, a digital CD output can be processed usingthe present invention, and the result can be recorded on ordinary stereoaudio tape. The present invention restores information that has beenlost during digital or analog processing, as well as during thetransduction of the original sound, and may be employed at a radio ortelevision broadcasting station to improve the quality of the audiosignal received by the listeners.

Further objectives, advantages and novel features of the invention willbecome apparent from the detailed description which follows.

BRIEF DESCRIPTIONS OF THE DRAWINGS

FIG. 1 is a block diagram of the entire system for increasing thebandwidth of recorded and live signals, adding harmonics and phaseinformation lost during the transduction and recording processes, andfor converting the result to duty-cyclemodulation and amplification;

FIG. 2 is a block diagram of the signal flow of the processor part ofthe system showing cross-overs that homogenize the Right and Left stereochannels and produce phase distributed monaural;

FIG. 3 displays the details of the processor portion of the system;

FIG. 4 shows the details of the analog-to-digital (A/D) converter andthe digital amplifier parts of the system;

FIG. 5A is the phase shift network used in the processor;

FIG. 5B illustrates the phase shift characteristics of the networkdisplayed in FIG. 5A;

FIG. 6A is the major high frequency harmonic producing petwork of theprocessor;

FIG. 6B illustrates how the network adds harmonics to the signals;

FIG. 7 illustrates how the processor separates harmonic frequencies intime as a function of frequency;

FIG. 8A shows actual oscilloscope trace of a single channel of a stereorecording;

FIG. 8B shows an actual oscilloscope trace for a single channel ofstereo that has been processed with the processor in FIG. 3;

FIG. 9A shows an actual oscilloscope trace of the two stereo channelssimultaneously input into an oscilloscope (the Right channel into the Xinput and the Left channel into the Y input);

FIG. 9B shows an actual oscilloscope trace for the signals processed bythe circuit of FIG. 3 and as described above under FIG. 9A;

FIG. 10A illustrates the output of a Compact Disc as a function of timeand frequency as taken from the screen of an audio spectrum analyzer;

FIG. 10B illustrates an actual trace from an audio spectrum analyzer forthe same passage shown in FIG. 10A, but processed with the circuit shownin FIG. 3;

FIG. 11A shows the output of the A/D converter for amplitude andfrequency and phase changes going in, frozen in time as a function offrequency;

FIG. 11B displays a typical duty-cycle-modulation as a time sequence(actual oscilloscope trace);

FIG. 12 shows an alternative embodiment for blocks number 20 and 21 inFIG. 3.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

With reference to the drawings, FIG. 1 shows the overall systemconsisting of the analog processor and the A/D converter and DigitalAmplifier. It is noted that the processor can function independently, ascan the converter and the digital amplifier. Indeed, the processor is aline device that can accept outputs from pre-amps, tape decks, CDplayers, or any source that could be plugged into any standard poweramplifier. Likewise, the A/D converter will accept line level inputs andcan be used to digitize any line level analog input. The digitalamplifier will accept any train of digital pulses and convert them toalternating current in order to drive transducer loads such as speakers,standing wave generators, tuned cavity resonators, hydrophones, etc.

FIG. 2 displays a block diagram of the analog processor by itself inorder to show the overall signal flow through the-circuit. The detailsof the processing is given in FIG. 3.

FIG. 3 displays Left and Right Channel inputs, with the Right Channelprocessing being identical to the Left Channel; therefor, only the LeftChannel processing will be described. Item 12 is an input buffer. Theoperational amplifier can be of any variety compatible with the pin-outof a 741, however, higher slew rate devices (>12 volts/microsecond) withJ or Bi-FET inputs are more desirable. Even video op-amps can be used.The buffer op-amp can be provided with a wide range of supply voltagesas specified by the manufacturer. However, in general, the higher thesupply voltages, the more dynamic range available for the audio.

The output of the unity gain input buffer (which also acts to reducerepetitive noise) is fed into phase shifter 13 and to the non-invertinginput of op-amp 2R Phase shifter 13's noninverting, inverting, andfeedback resistors (24, 25, 26) along with capacitor 32 are selected toprovide unity gain and a phase shift that delays higher frequencies morethan lower frequencies. The all-pass filter formed by the circuit of 13is described in more detail in FIG. 5. The Right Channel signal iscrossed over to the non-inverting input of op-amp 2L in 13. Note thatthis action does not cause a mining of the signal by way of a simpleaddition as formed in mixer circuits, rather, the Right Channel signalis fed into the non-inverting input of the Left Channel phase shifter(13) along with the phase shifted Left Channel information. What exitsop-amp 2L is a combination of Left Channel information, phase shiftedleft channel information, and (via the feedback resistor) a combinationof these signals. The effect is to provide complex phase informationthat was present at the live performance, but compressed during thetransduction and recording sequences. The output is fed into buffer 14,whose action is the same as buffer 12. The signal now goes into phaseshifter 15 whose action is the same as phase shifter 13 with oneimportant exception: the amount of phase shift is controlled bypotentiometer 28. The signal now proceeds to the harmonic enhancer 16.This circuit is discussed in detail in FIG. 6. After passing throughunity gain buffer 17, whose function is the same as 12 and 14, thesignal is summed in 18 with the sum of Left in and Right in. The sum ofLeft in and Right in is developed in 20 and 21. As shown, the twosignals are,summed in the inverting input to op-amp 10. All values ofresistors, 42, 43, 44, and 46, are the same, in order to produce nogain. However, the output of 10 is inverted. Therefore, it is runthrough inverting network 21. Resistors 45 and 47 are the same value;however, 48 is selected to provide 3dB gain. The reason for this is tocompensate for the 3dB gain produced by the cross-overs in 13 and 15(when 28 equals 10Kohms or higher). The output of 11 is summed with theprocessed Left Channel and the processed Right Channel in order torestore a “center” that is created on recordings where vocals are“panned to the center” but are canceled by the crossover process whichcan (at maximum) cause a 2π radian (360 degree) phase shift.

After the summation in 18, the signal goes through one more unity gainbuffer stage (19) where either the signal is attenuated 3 dB tocompensate for further increases in gain when 28 is near zero ohms, or Rin 19 is eliminated and 2R is made zero ohms and 22 is used to boost theRight and Left Channels in Bypass mode by 3 dB for reasons stated above.If circuit 22 is employed then R in 19 is not required and 2R=0. DPDTswitch 23 is used to switch between Process and Bypass. If theattenuation resistors in 19 are used, then Left in and Right in godirectly to the switch as shown in 23. The switch can have an indicator.

FIG. 4 shows the A/D converter, 100, and the digital amplifier 101.Referring to 100: the output of the processor in FIG. 3 goes into theinput of 100. R1 sets the input impedance and provides acharge/discharge path for C1. C1 blocks direct current from pin 2 of ICIfrom mixing with the input signal. IC1 is a regulating Pulse WidthModulator. However, as will be seen, in this case, the circuit willperform as a duty-cycle-modulator with a fixed carrier frequency. R2,R3, and R4 set the output duty cycle Q point, while R5 and C2 set theoscillation frequency. C3 and C4 filter the direct current input to IC1.R6 provides a current path for the base of Q1, a switching transistor,when it is “on”. R6 also pulls down the output of IC1. D1 keeps Q1 “off”when the output of IC1 (pins 11 and 15) are “high”. Q1 provides a fastdischarge of C5 through R9 when the output of IC1 is in the highimpedance (low logic) state. PS1 provides dc power to the chip, and itcan be any voltage/current recommended by the manufacturer of IC1.Referring to 101: C5 couples the output of IC1 with its dc bias to thepush-pull amplifier stage. It functions as an “anti-rectifier”, blockingdc. R9 provides a charge/discharge path for C5. Field Effect TransistorsQ2 and Q3 act as switches for positive and negative parts of the signal,shifting voltage, current and power to the requirements of the load.

D2 and R7 cause Q2 to turn on slightly slower than it turns off, so Q2and Q3 will never be on at the same time. D3 and R8 do same thing forQ3. PS2 and PS3 provide the plus and minus load power for the output.

FIG. 4 circuits perform the functions of converting the input from FIG.3 to a digital signal retaining all of the information. in the analogsignal. The instantaneous input amplitude values (which containfrequency and phase information) are sampled about 50,000 times/secondand converted into a corresponding duty-cycle-modulation of the digitaloutput. For zero volts amplitude in, the output duty cycle will beexactly 50% (adjusted so by R3 in 100). For any positive input value,the output duty cycle will be between 50% and 100%. For any negativeinput value the output duty cycle will be between zero and 50%. 100% isdefined as positive and zero % is defined as negative, i.e., inputamplitude changes are converted into output phase changes.

In order to drive significant loads (above 200 ma), the output of theconverter must go through a power buffer consisting of power FET's and adual power supply capable of driving the loads appropriately.

Referring to FIG. 5, the all pass circuit shown in 5A has thecharacteristics shown in 5B when R1 and C are “equal” (100K and 0.1microfarads). R2 and R3 are equal so that no net gain is caused,10Kohms, e.g.

FIGS. 6A and 6B show the harmonic enhancer of 16 in FIG. 3. R1 isselected to be about 10 times more resistance than R2, and R3 isselected to about 75 to 100 times more than R2. The result is that anyhigh frequency input signal to the enhancer network will developharmonics without experiencing a gain increase.

FIG. 7 displays how delayed high frequencies are actually separated inthe time domain, allowing the brain time to synthesize the informationinto a three dimensional hologram.

FIG. 8A shows an actual oscilloscope tracing of a passage from a CDrecording.

FIG. 8B shows the same passage after traversing the processor describedin FIG. 3.

FIG. 9A shows a typical, processed output from a CD where the Rightchannel was placed into the X input of an oscilloscope, and the Leftchannel was input to the Y input.

FIG. 9B shows the same passage Lissajous pattern after being processedby the circuitry of FIG. 3. Note that the signal now containsmulti-dimensional phase information extracted from the two-dimensionalstereo recording.

FIGS. 10A and 10B show before and after processing with the circuit ofFIG. 3. Note that virtually no harmonics were present beyond 20 KHzbefore processing; whereas, harmonics are evident out to the limit ofthe spectrum analyzer used—25 KHz—after processing.

FIG. 11A shows the output of 100 in FIG. 4, frozen in frequency. Thisindicates that all the necessary information contained in the analogsignal (amplitude, frequency, and phase) have been converted intoequivalent phase shifts of a duty cycle modulated pulse whose nominal(at zero input) duty cycle is 50%. FIG. 11B shows the signal frozen intime. Note that the frequency (time between I, II, and III) is all thesame; only the duty cycle has changed. The fall times of a, b, and chave changed, not the rise times.

FIG. 12 shows an alternative embodiment of the circuits in 20 and 21 inFIG. 3. If Left in and Right in are summed into the non-inverting inputof the op-amp, then R1 and R2 can be selected to provide 3 dB gain,thereby combining 21 and 22 in FIG. 3.

Having now fully set forth the preferred embodiments and certainmodifications of the concept underlying the present invention, variousother embodiments as well as certain variations and modifications of theembodiments herein shown and described will obviously occur to thoseskilled in the art upon becoming familiar with said underlying concept.It is to be understood, therefore, that the invention may be practicedotherwise than as specifically set forth herein.

What is claimed is:
 1. A signal processor comprising: (a) a firstchannel input terminal for inputting an audio signal; (b) a first unitygain stage buffer for receiving and buffering said input audio signal;(c) a first signal processing stage connected to said first unity gainstage buffer for processing said buffered audio signal by generatingharmonic frequencies related to fundamental frequencies in said signaland phase shifting said fundamental and harmonic frequencies as afunction of frequency, said phase shift increasing in a negativedirection with increasing frequency to cause higher frequency signals tolag lower frequency signals; (d) a second unity gain stage buffercoupled to said first signal processing stage for buffering theprocessed audio signal; (e) an output terminal coupled to said secondunity gain stage buffer for coupling the buffered and processed signalto an output device.
 2. A signal processor as claimed in claim 1,wherein said first signal processing stage further comprises: (a)operational amplifiers for said unity gain buffers and said harmonicgeneration and said phase shift stages; (b) an RC circuit forcontrolling said phase shifting function and said harmonic generation;(c) means for coupling said first channel to said second channel toprovide signal mixing between said channels by resistive means.
 3. Asignal processor as claimed in claim 2, wherein said means to couplesaid first and second channel further comprises: (a) first couplingmeans for coupling the first stage of said first channel to the secondstage of the said second channel, and (b) second coupling means forcoupling the second stage of said first channel to the first stage ofsaid second channel.
 4. A signal processor as claimed in claim 3,wherein said coupling means is repeated in the second processing stageof said cascade, and said means comprises resistances.
 5. A signalprocessor as claimed in claim 1, further comprising a second signalprocessing stage coupled to said first signal processing stage in acascaded configuration.
 6. A signal processor as claimed in claim 5,wherein said cascaded arrangement of first and second signal processingstages form a first of two channels, and further comprising a secondchannel comprising: (a) a second channel input terminal for receiving asignal; (b) a second unity gain stage buffer for receiving said inputsignal; (c) a second signal processing stage coupled to said inputterminal for processing said audio signal by generating harmonicfrequencies related to fundamental frequencies in said signal and phaseshifting said fundamental and harmonic frequencies as a function offrequency, said phase shift increasing in a negative direction withincreasing frequency to cause higher frequency signals to lag lowerfrequency signals; (d) a third unity gain stage buffer coupled to saidsecond signal processing stage; (e) a second output terminal coupled tosaid third unity gain stage buffer for coupling the processed signal toan output device.
 7. A signal processor as claimed in claim 6, furthercomprising a harmonic enhancing circuit in each channel.
 8. A signalprocessor as claimed in claim 7, wherein the two channels are combinedat the input terminals of said processors and the combined signal addedto each output channel prior to the final output buffer stage of eachchannel.
 9. A signal processor claimed in claim 6, further comprising:(a) means for bypassing said signal processing stages, (b) a switch forconnecting said input terminals to said output terminals, (c) a visualindicator for indicating the state of said switch.
 10. A signalprocessor as claimed in claim 6, further comprising: (a) a bypass switchfor connecting said input terminals to an amplifier whose output isconnected to the said signal processor's output terminals, (b) a visualindicator for indicating the state of said switch.
 11. A signalprocessor as claimed in claim 6, wherein the operational amplifier ineach of said harmonic generation and phase shift stages include afeedback resistance, said feedback resistance being variable in at leastone of said stages.
 12. A method for carrying out one and two stageprocessing of a first channel input signal and a second channel inputsignal, said method comprising the steps of: (a) generating harmonicfrequencies related to fundamental frequencies in each of said first andsecond channel input signals and adding said harmonic frequencies tosaid first and second channel input signals, respectively; (b) phaseshifting said fundamental and harmonic frequencies as a function offrequency in each of said first and second channel input signals, saidphase shift increasing in a negative direction with increasing frequencyto cause higher frequency signals to lag lower frequency signals, saidharmonic generating and phase shifting steps thereby producing apartially processed first channel signal and a partially processedsecond channel signal; (c) generating harmonic frequencies related tofundamental frequencies in each of said partially processed firstchannel and second channel signals and adding said harmonic frequenciesto said partially processed first and second channel signals,respectively; (d) phase shifting said fundamental and harmonicfrequencies of each of the partially processed first and second channelsignals as a function of frequency, said phase shift increasing in anegative direction with increasing frequency to cause higher frequencysignals to lag lower frequency signals, said harmonic generating andphase shifting steps thereby producing a fully processed first channelsignal and a fully processed second channel signal; (d) mixing at leasta portion of said first channel input signal with a fully processed orpartially processed second channel input signal, and mixing at least aportion of said second channel input signal with a fully processed orpartially processed first channel input signal; (e) controlling theamount of harmonic frequency generation and phase shifting that iscarried out on at least one of said input signals and said partiallyprocessed signals.